Driving device

ABSTRACT

A driving device comprises a first transistor (B 13 ), a second transistor (B 14 ), and a resistance element. The first transistor (B 13 ) has one terminal receiving a pulsed current and a control terminal connected to the one terminal. The second transistor (B 14 ) has one terminal connected to at least one load, the other terminal connected to a reference potential together with the other terminal of the first transistor (B 13 ), and a control terminal connected to the control terminal of the first transistor (B 13 ). The resistance element is connected between the control terminal of the first transistor (B 13 ) and the other terminal of the first transistor (B 13 ).

TECHNICAL FIELD

The present invention relates to a driving device that performs drivingcontrol of a load (light emitting diode and the like).

BACKGROUND ART

As a backlight for a LCD (Liquid Crystal Display) panel (e.g., carnavigation monitor), currently, a cold cathode fluorescent lamp (CCFL)is chiefly used; however, from viewpoints of an Hg-free movement andadvantages in high brightness, energy saving, longevity and the like, inrecent years, a white LED (Light Emitting Diode) is in practical use;and various technologies are disclosed and proposed for a LED drivingdevice (so-called LED driver) that performs drive control (e.g., see apatent document 1).

Patent document: JP-A-2007-13183

DISCLOSURE OF THE INVENTION Problems to be Solved by the Invention

When making a LED emit light under PWM control, a controllableduty-ratio range is limited by characteristics of the element.

Besides, because of the characteristics of the element, accuracy of anelectric current to make the LED emit light is low.

In light of the above problems, it is an object of the present inventionto provide a driving device that is able to extend the controllableduty-ratio range and raise the accuracy of a control current regardlessof a change in the element.

Means for Solving the Problem

To achieve the above object, a driving device according to the presentinvention is so structured (first structure) as to include: a firsttransistor into one terminal of which a pulse-shape current is input andthe one terminal of which is connected with a control terminal; a secondtransistor with one terminal of which at least one load is connected andthe other terminal of which is connected with a reference potentialtogether with the other terminal of the first transistor, and a controlterminal of which is connected with a control terminal of the firsttransistor; wherein a resistor element is connected between the controlterminal of the first transistor and the other terminal of the firsttransistor.

Besides, the driving device having the above first structure may be sostructured (second structure) as to further include: a second referencevoltage supply portion and a second reference voltage supply portionthat separately supply a predetermined voltage to one terminal of thefirst transistor and one terminal of the second transistor.

ADVANTAGES OF THE INVENTION

In the driving device according to the present invention, by inserting aresistor, it is possible to speed up rising of the transistor, so thatit becomes possible to extend a range of controllable duty ratios.

Besides, in the driving device according to the present invention, bymaintaining one terminal of the transistor at a constant voltage, itbecomes possible to raise the accuracy of a control current regardlessof a change in the element.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram showing a first embodiment of a semiconductordevice according to the present invention.

FIG. 2 is a pin arrangement diagram of a semiconductor device 10.

FIG. 3 is a table representing pin numbers, terminal names and functionsof external terminals.

FIG. 4 is a diagram for describing external connections of thesemiconductor device 10.

FIG. 5 is a setting table showing examples of constants of externalelements.

FIG. 6 is an input/output equivalent circuit diagram of externalterminals.

FIG. 7 is a table showing electric characteristics of the semiconductordevice 10.

FIG. 8 is a diagram showing an output stage of a FAIL1 signal.

FIG. 9 is a diagram showing an output stage of a FAIL2 signal.

FIG. 10 is a diagram for describing an open/short detection operation.

FIG. 11 is a truth table showing a correlation between input logics ofLED enable signals LEDEN1, LEDEN2 and on/off states of LED outputterminals LED1 to LED4.

FIG. 12 is a circuit diagram showing structural examples of a currentset portion 116 and a constant-current driver 117.

FIG. 13 is a graph showing a correlation between a resistor RSET and anoutput current ILED.

FIG. 14A is a timing chart showing an example of PWM light control(PWM=150 Hz, Duty=0.38%).

FIG. 14B is a timing chart showing an example of PWM light control(PWM=150 Hz, Duty=50%).

FIG. 14C is a liming chart showing an example of PWM light control(PWM=20 KHz, Duty=50%).

FIG. 15 is a graph showing a correlation between a resistor RT and anoscillation frequency FOSC.

FIG. 16 is a diagram for describing a connection relationship amongexternal elements related to operation of an OCP portion 111.

FIG. 17A is a waveform diagram for describing selection of a coil L1

FIG. 17B is a circuit diagram for describing selection of the coil L1.

FIG. 18 is a diagram for describing selection of an output capacitorCVOUT.

FIG. 19 is a diagram for describing selection of an input capacitorCVCC.

FIG. 20 is a diagram for describing selection of a load-switchtransistor Q1 and its soft start.

FIG. 21 is a graph showing a correlation between a gate-source capacityof the transistor Q1 and a soft start time.

FIG. 22 is a circuit diagram showing an LC resonance circuit of a DC/DCconverter.

FIG. 23 is a circuit diagram showing a first example (an ESR componentof an output capacitor) of a phase advance means.

FIG. 24 is a circuit diagram showing a second example (a CR componentconnected with a COMP terminal) of a phase advance means.

FIG. 25 is a timing chart showing an operation sequence of thesemiconductor device 10.

FIG. 26 is a block diagram showing a second embodiment of asemiconductor device according to the present invention.

FIG. 27 is a pin arrangement diagram of a semiconductor device 20.

FIG. 28 is a table representing pin numbers, terminal names andfunctions of external terminals.

FIG. 29 is a table representing electric characteristics of thesemiconductor device 20.

FIG. 30 is a circuit diagram showing structural examples of a currentset portion 220 and a constant-current driver 221.

FIG. 31A is a schematic diagram (in a case of a mirror ratio 1:1650)showing a correlation between a mirror ratio and a transistor size.

FIG. 31B is a schematic diagram (in a case of a mirror ratio 1:100)showing a correlation between a mirror ratio and a transistor size.

FIG. 32 is a layout diagram of zigzag-arranged resistors.

LIST OF REFERENCE SYMBOLS

-   -   10 semiconductor device    -   101 reference voltage generation portion (VREG portion)    -   102 switch    -   103 reduced-voltage protection portion (UVLO portion)    -   104 temperature protection portion (TSD portion)    -   105 overvoltage protection portion (OVP portion)    -   106 input buffer    -   107 oscillator portion    -   108 PWM comparator    -   109 control logic portion    -   110 driver    -   111 overcurrent protection portion (OCP portion)    -   112 comparator    -   113 error amplifier    -   114 soft start portion    -   115 input buffer    -   116 current set portion    -   117 constant-current driver    -   118 open/short detection portion    -   119 input buffer    -   120 input buffer    -   20 semiconductor device    -   201 reference voltage generation portion (VREG portion)    -   202 reduced-voltage protection portion (UVLO portion)    -   203 temperature protection portion (TSD portion)    -   204 short protection portion (SCP portion)    -   205 overvoltage protection portion (OVP portion)    -   206 overcurrent protection portion (OCP portion)    -   207 comparator    -   208 control logic portion    -   209 input buffer    -   210 oscillator portion    -   211 slope-voltage generation portion    -   212 PWM comparator    -   213 driver control portion    -   214 driver    -   215 N-channel type field effect transistor    -   216 driver    -   217 error amplifier    -   218 soft start portion    -   219 input buffer    -   220 current set portion    -   221 constant-current driver    -   222 open/short detection portion    -   223 input buffer    -   224 input buffer

BEST MODE FOR CARRYING OUT THE INVENTION

FIG. 1 is a block diagram showing a first embodiment of a semiconductordevice according to the present invention.

First, an overview of a semiconductor device 10 according to the presentembodiment is described.

The semiconductor device 10 is a 36 V-resistant white-LED driver IC; anda voltage step-up DC/DC converter and a four-channel outputconstant-current driver are integrated into one chip. The semiconductordevice 10 is able to perform light control of the white LED by using anyof PWM [Pulse Width Modulation] control and VDAC control.

Next, features of the semiconductor device 10 according to the presentembodiment are described.

A first feature is that the input voltage range of a power-supplyvoltage VCC is 4.5 to 30[V]. A second feature is that a voltage step-upDC/DC converter is incorporated. A third feature is that a four-channelconstant-current driver for supplying an output current ILED to a LED(the maximum electric-current value: 150 [mA]). A fourth feature is thatthe semiconductor device 10 interacts with PWM light control (the dutyratio: 0.38 to 99.5 [%]). A fifth feature is that various protectionfunctions (UVLO [Under Voltage Lock Out], OVP [Over Voltage Protection],TSD [Thermal Shut Down], OCP [Over Current Protection]) areincorporated. A sixth feature is that a detection function for detectinga LED abnormal state (open/short) is incorporated. A seventh feature isthat an HSOP-M28 package (see FIG. 2) is employed.

The semiconductor device 10 according to the present embodiment is usedfor drive control of a backlight of a car navigation monitor, backlightsof medium- and small-sized LCD panels and the like.

The semiconductor device 10 having the above features according to thepresent embodiment, as shown in FIG. 1, is composed of an integrationof: a reference voltage generation portion 101 (hereinafter, called aVREG portion 101); a switch 102; a reduced-voltage protection portion103 (hereinafter, called a UVLO portion 103); a temperature protectionportion 104 (hereinafter, called a TSD portion 104); an overvoltageprotection portion 105 (hereinafter, called an OVP portion 105); aninput buffer 106; an oscillator portion 107; a PWM comparator 108; acontrol logic portion 109; a driver 110; an overcurrent protectionportion 111 (hereinafter, called an OCP portion 111); a comparator 112;an error amplifier 113; a soft start portion 114; an input buffer 115; acurrent set portion 116; and a constant-current driver 117; anopen/short detection portion 118; and input buffers 119 and 120.

Here, it is possible to roughly divide the above circuit portion of thesemiconductor device 10 into four blocks of: a VREG block (the VREGportion 101); a voltage step-up DC/DC controller block (the switch 102,input buffer 106, oscillator portion 107, PWM comparator 108, controllogic portion 109, driver 110, OCP portion 111, comparator 112, erroramplifier 113 and soft start portion 114); a current driver block (theinput buffer 115, current set portion 116, constant-current driver 117,open/short detection portion 118 and input buffers 119 and 120); aprotection block (the UVLO portion 103, TSD portion 104, OVP portion105).

Besides, the semiconductor device 10 according to the present embodimentincludes 28 external terminals (1st to 28th pins) as means for securingelectric connections with outside.

FIG. 2 is a pin arrangement diagram of the semiconductor device 10 andFIG. 3 is a table which shows pin numbers, terminal names and functionsof the external terminals. In FIG. 2, wide terminals disposed at bothsides of central portions of the semiconductor device 10 are FINterminals that are linked to subground and improve heat radiation.

Next, detailed description of external connections of the semiconductordevice 10 is performed.

FIG. 4 is a diagram for describing the external connections of thesemiconductor device 10.

As for external elements shown in FIG. 4, it is desirable thatdecoupling capacitors CVCC, GREG are connected as close to IC pins aspossible.

Because a large current is likely to flow in a CS terminal (22nd pin), aGND terminal (7th pin) and a PGND (21st pin), it is desirable toseparately wire them and lower the impedances.

It is necessary to make sure that noise does not appear on a VDACterminal (8th pin), an ISET terminal (9th pin), a RT terminal (26th pin)and a COMP terminal (28th pin).

It is necessary to make sure that a PWM terminal (5th pin), a SYNCterminal (6th pin), a LED1 terminal (12th pin), a LED2 terminal (14thpin), a LED3 terminal (15th pin) and a LED4 terminal (17th pin) do notinfluence patterns around them because they are switched.

It is desirable that thick-line portions in FIG. 4 are designed withwide patterns and a layout as short as possible.

Here, in the semiconductor 10 according to the present embodiment,because a power transistor Q2 is externally connected, it becomespossible to raise heat radiation.

FIG. 5 is a setting table showing examples of constants of externalelements. Here, the constants shown as examples in this figure areconstants whose operations are confirmed at the power-supply voltageVCC=12[V], LED 5 in series and 4 in parallel, the output current ILED=50[mA]. Accordingly, because the optimum values are different depending onuse conditions and the like, it is desirable to decide on the constantsafter a sufficient evaluation.

FIG. 6 is an input/output equivalent circuit diagram of the externalterminals.

As shown in FIG. 6, electrostatic protection diodes are connected withall the external terminals of the semiconductor device 10.

Besides, as for the PWM terminal (5th pin), SYNC terminal (6th pin),VDAC terminal (8th pin), ISET terminal (9th pin), LEDEN1 terminal (10thpin), LEDEN2 terminal (11th pin), CS terminal (22nd pin), SWOUT terminal(23rd pin), EN terminal (24th pin), OVP terminal (25th pin), RT terminal(26th pin), SS terminal (27th pin), COMP terminal (28th pin) that arecontrol related terminals, a structure is employed, in which thecathodes of electrostatic diodes on the upper sides (on a route sidewhere electric charges are pulled out from a signal line to apower-supply line) are not connected with application terminals of thereference voltage VREG and the power-supply voltage VCC but with anapplication terminal of an intermediate voltage CL10V (e.g, 10[V]; seethe most right below of FIG. 6).

According to such structure, in a case where the power-supply voltageVCC is not applied, or in a case where the reference voltage VREG is notgenerated by an enable signal EN, even if a positive voltage is appliedto an external terminal, an overcurrent does not flow in the referencevoltage line and the power-supply voltage line via an electrostaticprotection diode, accordingly, it becomes possible to protect breakdownand erroneous operation of the circuit.

FIG. 7 is a table showing electric characteristics of the semiconductordevice 10 that has the above structure. Here, the electriccharacteristics shown in FIG. 7 represent numerical values at thepower-supply voltage VCC=12 [V], ambient temperature Ta=25[° C.] unlessotherwise specified.

Next, detailed description of the VREG block (VREG portion 101) of thesemiconductor device 10 is performed with reference to the above FIG. 1and the like.

The VREG portion 101 is a means that generates the reference voltageVREG (5[V] (Typ.)) from the power-supply voltage VCC (12[V]) input intothe VCC terminal (1st pin) when the enable signal EN input into the ENterminal (24th pin) is in a high level. This reference voltage VREG isused as a power supply for an internal circuit and also used to fix aterminal at a high-level voltage outside the IC.

Besides, the VREG portion 101 includes a UVLO function, begins operationat 2.9[V] (Typ.) or higher, and stops the operation at 2.8[V] (Typ.) orlower.

Here, the VREG terminal (4th pin) is an external terminal to connect thecapacitance CREG (10 μF (Typ.)) for phase compensation. By connectingsuch capacitance CREG for phase compensation, it becomes possible tostabilize circuit operation of the VREG portion 101.

Next, detailed description of a self-diagnosis function of thesemiconductor device 10 is performed.

The semiconductor device 10 according to the present embodiment, torepresent an operation state of a protection circuit incorporated initself, includes a function to output the FAIL1 signal and the FAIL2signal in an open-drain fashion from the FAIL1 terminal (3rd pin) andthe FAIL2 terminal (20th pin), respectively.

If any of the UVLO portion 103, TSD portion 104, OVP portion 105 and OCPportion 111 detects an abnormal state and the output signal is broughtto a low level, the control logic portion 109 brings the FAIL1 signal tothe low level via an output stage shown in FIG. 8 and fixes the SWOUTterminal (23rd pin) at the low level, thereby stopping the voltagestep-up operation.

However, because the OCP portion 111 is of a pulse-by-pulse type, afterthe SWOUT terminal is fixed at the low level for only one period decidedon by the oscillation frequency FOSC of the voltage step-up DC/DCconverter, the voltage step-up operation is resumed. According to suchpulse-by-pulse type, because it is possible to limit a current withoutcompletely stopping the circuit operation, automatic resumption isperformed with no delay even if the circuit is stopped by erroneousoperation, so that it is easy for the user to operate.

Besides, if at least one of the UVLO portion 103, TSD portion 104 andOVP portion 105 detects an abnormal state, all the LED1 terminal, LED2terminal, LED3 terminal and LED4 terminal (12th pin, 14th pin, 15th pin,17th pin) are opened (high impedance).

Besides, the FAIL1 signal output from the FAIL1 terminal (3rd pin) andthe LOADSW signal output from the LOADSW terminal (2nd pin) are signalsinverted from each other; if the FAIL1 signal is brought to the lowlevel, the LOADSW signal is brought to the high level by means of theswitch 102. Accordingly, if any of the UVLO portion 103, TSD portion104, OVP portion 105 and OCP portion 111 detects an abnormal state, aload switch (the P-channel type field effect transistor Q1 in FIG. 4)that is externally connected with the LOADSW terminal (2nd pin) isturned off. Accordingly, in an abnormal time of the semiconductor device10, the voltage step-up operation is stopped, so that it becomespossible to prevent the IC from being broken, emitting smoke, orburning.

On the other hand, the FAIL2 signal output from the FAIL2 terminal (20thpin) is brought to the low-level output via an output stage shown inFIG. 9 if the open/short detection portion 118 detects an abnormal state(an open state or a short state). Here, the FAIL2 signal output from theopen/short detection portion 118 is of a latch type, and release of thelatch is performed based on on/off (and on/off of the UVLO signal) ofthe enable signal EN.

As shown in FIG. 10, if LED terminal voltages V1 to V4 (the respectiveterminal voltages of the LED1 terminal to LED4 terminal) that are to bemaintained at a predetermined LED control voltage VLED (0.8 [V] (Typ.))become 0.15 [V] (Typ.) or lower, the open/short detection portion 118determines that the LED terminal is opened; further, if a terminalvoltage VP (a divided voltage of an output voltage VOUT) of the OVPterminal (25th pin) reaches 1.7 [V] (Typ.), the open/short detectionportion 118 transmits an instruction to the constant-current drive 117so as to turn off a current output for the LED terminal that is judgedto be opened and shifts the FAIL2 signal to the low level. Here, in theexample in FIG. 10, a case where the LED1 terminal is opened is shown asan example.

As described above, by performing the open detection of the LED terminaland the off control of the electric-current output in a two-stepfashion, it becomes possible to avoid an unnecessary shutdown.

Here, as for the above open detection, it is possible to substituteovervoltage detection by the OVP portion 105. Specifically, in the OVPportion 105, it is detected that the terminal voltage VP of the OVPterminal reaches a predetermined overvoltage detection voltage VDOVP(2.0 [V] (Typ.)); the voltage step-up operation of the DC/DC converteris stopped; and the electric-current outputs of all the channels areturned off; accordingly, without performing the open detection, it ispossible to turn off the electric-current outputs of all the channels byperforming the overvoltage detection that doubles as the open detection.

Here, to turn off only a channel that is opened, as described above, itis sufficient to identify the LED terminal that is opened and turn offonly the channel by monitoring the LED terminal voltages V1 to V4.Especially, as for adaptation to an application (a backlight drivedevice for a car navigation monitor and the like) in which troubleoccurs in the use if the electric-current outputs of all the channelsare turned off, the structure according to the present embodiment thatis able to separately turn off the electric-current outputs of therespective channels is desirable.

Besides, the open/short detection portion 118 determines that a shortoccurs if the LED terminal voltages V1 to V4 become 4.5 [V] (Typ.) orhigher. In other words, if a difference between the LED terminalvoltages in a normal time and an abnormal time becomes 3.7 [V] (=4.5[V]−0.8 [V]) (Typ.) or higher, the short is detected.

Here, because a forward-direction drop voltage VF of the white LED isabout 3.4 [V], in the above setting example, a short is not detectedeven if only one LED shorts; but a short is detected if two or more LEDsshort. According to the setting of such a threshold level, it becomespossible to avoid an unnecessary shutdown within an extent where a LEDshort occurs but serious trouble is not caused in the use.

As described above, the short detection in the open/short detectionportion 118 means a detection operation in which for example, one LEDthat constitutes any one of LED trains that are connected separately andexternally with the LED1 terminal to LED4 terminal goes to a short state(a short-circuit state between the anode and the cathode); as a resultof this, a forward-direction drop voltage of the entire LED trainbecomes low by the forward-direction drop voltage VF of the LED thatgoes to the short state, so that a state in which one LED terminalvoltage becomes higher than the other LED terminal voltages by theforward-direction drop voltage VF of the LED is detected. Accordingly,as shown in FIG. 1, the open/short detection portion 118 and the OVPportion 105 are formed as protection blocks separate from each other.

Here, in the semiconductor device 10 according to the presentembodiment, if an open/short is detected, thereafter, the shortdetection signal is masked. Describing with reference to the example inFIG. 10, after an open of the LED1 terminal is detected, the shortdetection signals of the other LED2 terminal to LED4 terminal aremasked. According to such mask control, as a result of the fact that theLED1 terminal is opened, even in a case where the LED terminal voltageV1 drops almost to the GND, in response to this, the output voltage VOUTrises; by following this, the LED terminal voltages V2 to V4 rise higherthan usual, this is not erroneously detected as a short. Here, the opendetection signal is not masked even after the open/short detection.

Besides, the above short detection signal is also masked in an off timeof the output current ILED by the PWM drive. According to such maskcontrol, even if the LED terminal voltages V1 to V4 leap in an off timeof the output current ILED, this is not erroneously detected as a short.As for the above mask control, considering that a delay between thelogic-shift timing of the PWM signal and the on/off timing of the outputcurrent ILED occurs, it is sufficient to mask from the timing ofstarting to flow the output current ILED (the timing the outputtransistor of the constant-current driver 117 is turned on) to thetiming the PWM signal falls to the low level (see FIG. 13 laterdescribed).

Besides, if additional capacitance is connected with the LED1 terminalto LED4 terminal, the LED terminal voltages V1 to V4 become slow to dropand the short detection can malfunction; accordingly, it is necessary totake care. Besides, because both FAIL1 signal and FAIL2 signal are ofthe open-drain type, the FAIL1 terminal and the FAIL2 terminal arepulled up to the application terminal of the reference voltage VREG viaresistors (resistors RFL1, RFL2 in FIG. 4).

Next, detailed description of the current driver block (the input buffer115, current set portion 116, constant-current driver 117, open/shortdetection portion 118 and input buffers 119 and 120) of thesemiconductor device 10 is performed.

Of the LED output terminals LED1 to LED4, if there is an output terminal(and a train of LEDs that do not go on) that does not use the outputcurrent ILED from the constant-current driver 117, it is possible toseparately turn off the electric-current outputs for the LED outputterminals LED1 to LED4 by using the LEDEN1 terminal (10th pin) and theLEDEN2 terminal (11th pin).

FIG. 11 is a truth table showing a correlation between input logics ofLED enable signals LEDEN1, LEDEN2 and on/off states of the LED outputterminals LED1 to LED4.

Here, if a LED terminal that is not used is opened without using the LEDenable signals LEDEN1, LEDEN2, the open detection malfunctions in theopen/short detection portion 118. Besides, even if the electric-currentoutput for the LED terminal is turned off by using the LED enablesignals LEDEN1, LEDEN2, the input stage of the error amplifier 113operates; accordingly, it is desirable that the LED1 terminal to LED4terminal are not fixed to the GND but opened or connected with theapplication terminal of the constant voltage VREG. Besides, when theoutput current ILED is PWM-driven, it is desirable not to switch the LEDenable signals LEDEN1, LEDEN2.

Next, a method for setting the output current ILED is described indetail with reference to FIG. 12.

FIG. 12 is a circuit diagram showing structural examples of the currentset portion 116 and the constant-current driver 117.

As shown in FIG. 12, the current set portion 116 includes: anoperational amplifier A1; a direct-current voltage source A2; annpn-type bipolar transistor A3; resistors A4, A5; pnp-type bipolartransistors A6 to A9; and a resistor A10 (the resistance value R).

A first non-inverting input terminal (+) of the operational amplifier A1is connected with the VDAC terminal (8th pin). A second non-invertinginput terminal (+) of the operational amplifier A2 is connected with apositive-polar terminal of the direct-current voltage source A2, and apredetermined constant voltage VISET (=2.0 [V]) is applied. Anegative-polar terminal of the direct-current voltage source A2 isconnected with a ground terminal. An inverting input terminal (−) of theoperational amplifier A2 is connected with the ISET terminal (9th pin).A base of the transistor A3 is connected with an output terminal of theoperational amplifier A1. An emitter of the transistor A3 is connectedwith the ISET terminal.

One terminal of each of the resistors A4, A5 is connected with theapplication terminal of the reference voltage VREG. The other terminalof the resistor A4 is connected with an emitter of the transistor A6.The other terminal of the resistor R5 is connected with an emitter ofthe transistor A7. Bases of the transistors A6, A7 are connected witheach other and the connection node is connected with a collector of thetransistor A7. A collector of the transistor A6 is connected with anemitter of the transistor A8. The collector of the transistor A7 isconnected with an emitter of the transistor A9. Bases of the transistorsA8, A9 are connected with each other and the connection node isconnected with a collector of the transistor A8. The collector of thetransistor A8 is connected with the collector of the transistor A3. Acollector of the transistor A9 is connected with the ground terminal viathe resistor A10.

On the other hand, as shown in FIG. 12, the constant-current driver 117includes 4 channels of output stages Ch1, Ch2, Ch3 and Ch4 that supplythe output current ILED to the LED1 terminal to LED4 terminal,respectively. Here, the output stage Ch1 includes: an operationalamplifier B1; an N-channel type field effect transistor B2; a resistorB3 (the resistance value 5R); a current mirror circuit B4 (the mirrorratio 1:1); a resistor B5 (the resistance value 5R); an operationalamplifier B6; an N-channel type field effect transistor B7; an resistorB8 (the resistance value 5R); a current mirror circuit B9 (the mirrorratio 1:10); an operational amplifier B10; a direct-current voltagesource B11; N-channel type field effect transistors B12 to B14; anoperational amplifier B15; a direct-current voltage source B16; anN-channel type field effect transistor B17; and a resistor B18.

A non-inverting input terminal (+) of the operational amplifier B1 isconnected the connection node of the transistor A9 and the resistor A10.An inverting input terminal (−) of the operational amplifier B1 isconnected with one terminal of the resistor B3. The other terminal ofthe resistor B3 is connected with the ground terminal. A drain of thetransistor B2 is connected with an input terminal of the current mirrorcircuit B4. A source of the transistor B2 is connected with one terminalof the resistor B3. A gate of the transistor B2 is connected with anoutput terminal of the operational amplifier B1. A power-supply inputterminal of the current mirror circuit B4 is connected with theapplication terminal of the reference voltage VREG.

A non-inverting input terminal (+) of the operational amplifier 136 isconnected with an output terminal of the current mirror circuit B4 andwith one terminal of the resistor B5. An inverting input terminal of theoperational amplifier B6 is connected with one terminal of the resistorB8. Both of the other terminals of the resistors B5, B8 are connectedwith the ground terminal. A drain of the transistor B7 is connected withan input terminal of the current mirror circuit 139. A source of thetransistor B7 is connected with one terminal of the resistor B8. A gateof the transistor B7 is connected with an output terminal of theoperational amplifier B6. A power-supply input terminal of the currentmirror circuit B9 is connected with the application terminal of thereference voltage VREG.

A non-inverting input terminal (+) of the operational amplifier B10 isconnected with a positive-polar terminal of the direct-current voltagesource B11. A negative-polar terminal of the direct-current voltagesource B11 is connected with the ground terminal. A drain of thetransistor B12 is connected with an output terminal of the currentmirror circuit B9. A source of the transistor B12 is connected with aninverting input terminal (−) of the operational amplifier B10. A gate ofthe transistor B12 is connected with an output terminal of theoperational amplifier B10.

A drain of the transistor B13 is connected with the source of thetransistor B12. Gates of the transistors B13, B14 are connected witheach other and the connection node is connected with the drain of thetransistor B12 and also connected with the ground terminal via theresistor B18. Both sources of the transistors B13, B14 are connectedwith the ground terminal.

A non-inverting input terminal (+) of the operational amplifier B15 isconnected with a positive-polar terminal of the direct-current voltagesource B16. A negative-polar terminal of the direct-current voltagesource B16 is connected with the ground terminal. A drain of thetransistor B17 is connected with the LED1 terminal. A source of thetransistor B17 is connected with an inverting input terminal (−) of theoperational amplifier B15 and also connected with a drain of thetransistor B14. A gate of the transistor B17 is connected with an outputterminal of the operational amplifier B15.

Here, because the other output stages Ch2 to Ch4 that constitute theconstant-current driver 117 include the same structure as that of theabove output stage Ch1, detailed description of them is skipped.

In the current set portion 116 and the constant-current driver 117 thathave the above structures, the output current ILED is set based on thefollowing formula (1).

ILED[mA]=min{VDAC,2.0[V]}/RSET[kΩ]×3300  (1)

In the above formula (1), a parameter min {VDAC, 2.0 [V])} is a voltagevalue that is the lower of the control voltage VDAC input into the VDACterminal (8th pin) and the constant voltage VISET (=2.0 [V])predetermined in the current set portion 116. Besides, a parameter RSETis a resistance value of the resistor RSET that is externally connectedwith the ISET terminal (9th pin); and a parameter 3300 (Typ.) is aconstant that is decided on in the constant-current driver 117.

Specifically, the resistor RSET is pulldowm-connected with the ISETterminal (9th pin), so that an electric current predetermined-gain times(e.g., 3300 times) larger than the reference current ISET flowing inthis is set as the maximum value (e.g., 50 [mA]) of the output currentILED.

Describing with reference to the example in FIG. 12, in theconstant-current driver 117, first, by using the operational amplifierB1, transistor B2 and resistor B3 (the resistance value 5R), a terminalvoltage Va (=ISET×R) of the resistor A10 is voltage/current-converted togenerate an intermediate current Ia (=1/5ISET) that is ⅕ the referencecurrent ISET. Next, by using the current mirror circuit B4, theintermediate current Ia is mirrored at 1:1 to generate an intermediatecurrent Ib (=1/5ISET). Next, by using the resistor B5 (the resistancevalue 5R), the intermediate current Ib is current/voltage-converted togenerate a terminal voltage Vb (=ISET×R). Next, by using the operationalamplifier B6, transistor B7 and resistor B8 (the resistance value 5R),the terminal voltage Vb of the resistor B5 is voltage/current-convertedto generate an intermediate current Ic (=1/5ISET). Next, by using thecurrent mirror circuit B9, the intermediate current Ic is mirrored at1:10 to generate an intermediate current Id (=2ISET) that is two timesas large as the reference current ISET. And, finally, by using thecurrent mirror circuit that includes the transistors B13, B14, theintermediate current Id is mirrored at 1:1650 to generate the outputcurrent ILED (=3300ISET) that is 3300 times as large as the referencecurrent ISET.

Here, to raise accuracy of the output current ILED, in the last-stagecurrent mirror circuit, by using the operational amplifiers B10, B15,drain-source voltages of the transistors B13, B14 are made identical toeach other (e.g., 0.3 [V]). Besides, the constant-current driver 117 isso structured as to generate the desired output current ILED byrepeating the voltage/current conversion and the current/voltageconversion based on the input reference current ISET. Accordingly, thenumber of resistor elements (the resistors B3, B5 and B8 in the examplein FIG. 12) used for the above conversion processes increases andtrimming chances increase. As described above, according to thestructure including many resistors that are able to be trimmed, byfinely adjusting the resistance values, it is possible to achieve arelative uneven width of ±4% and an absolute uneven width of ±6%, whichis able to contribute to reduction in the brightness unevenness and tolongevity of the LED.

FIG. 13 is a graph showing a correlation between the resistor RSET andthe output current ILED. Here, it is desirable to use a resistor having300 [k Ω] or smaller as the resistor RSET.

Besides, in a case where variable control (light control of the LED) ofthe output current ILED is performed by using the above control voltageVDAC, it is sufficient to set the input range at a range of 0.1 to 2.0[V]. By applying such control voltage VDAC, it becomes possible todecrease the output current ILED from the maximum value.

On the other hand, in a case where 2.0 [V] or higher is input as thecontrol voltage VDAC, as given by the above formula (1), the voltagevalue of the constant voltage VISET is selected; accordingly, the lightcontrol function by the control voltage VDAC is not used. Here, in acase where the light control by the control voltage VDAC is not used,from the viewpoint of avoidance of malfunction, it is sufficient not toopen the VDAC terminal but connect it with the application terminal ofthe reference voltage VREG (5 [V]).

In addition, in the semiconductor device 10 according the presentembodiment, besides the light control of the LED that uses the abovecontrol voltage VDAC, by using the PWM signal input into the PWMterminal (5th pin), the on/off control of the reference current ISET isperformed, so that it is also possible to perform the light control ofthe LED.

Specifically, based on the PWM signal, if a pulse current is generatedas the reference current ISET that serves as the reference for theoutput current ILED, the duty ratio of the PWM signal becomes the dutyratio of the output current ILED; accordingly, it becomes possible toseemingly decrease the output current ILED from the maximum value (or acurrent value decided on by the control voltage VDAC). Here, it issufficient to dispose an on/off control means (a pulse currentgeneration means) for the reference current ISET based on the PWM signalin the output stage (the previous stage of the constant-current driver117) of the current set portion 116.

Besides, in the semiconductor device 10 according to the presentembodiment, to raise response of the output current MED to the PWMsignal, a pull-down resistor B18 (500 [kΩ]) is inserted between the gateand the source of each of the transistors B13, B14 in the last-stagecurrent mirror circuit. According to such insertion of the pull-downresistor B18, because it is possible to speed up the rising of thetransistors B13, B14, it becomes possible to achieve increase (at theminimum duty ratio of 0.38[%] (150 [Hz])) in the PWM light controlcapability.

On the other hand, in a case where the light control by the PWM signalis not used (the duty ratio is 100%), it is sufficient to fix the PWMterminal at the high level (e.g., the constant voltage VREG). Here, itis desirable to insert a low pass filter (a cut-off frequency of 30[kHz]) into the PWM terminal.

FIGS. 14A to 14C are timing charts showing examples of the PWM lightcontrol, and each show a correlation between the PWM signal and theoutput current ILED. Here, FIG. 14A shows a case where the PWM signalhas a frequency of 150 [Hz] and a duty ratio of 0.38[%]; FIG. 14B showsa case where the PWM signal has a frequency of 150 [Hz] and a duty ratioof 50[%]. And, FIG. 14C shows a case where the PWM signal has afrequency of 20 [kHz] and a duty ratio of 50[%]. Here, although all thehorizontal axes of FIGS. 14A, 14B and 14 c are each a time axis, thefrequencies of the PWM signal are extremely different from each other;accordingly, the representing ranges are different from each other.Usually, the frequency of the PWM signal is set fixedly at about 100 to200 [Hz].

Next, a voltage step-up DC/DC controller block of the semiconductordevice 10 (a circuit block that includes: the input buffer 106;oscillator portion 107; PWM comparator 108; control logic portion 109;driver 110; OCP portion 111; comparator 112; error amplifier 113 andsoft start portion 114) is described in detail.

First, basic operation (voltage step-up operation) of the voltagestep-up DC/DC controller block is described in detail with reference tothe above FIGS. 1 and 4.

The transistor Q2 is an N-channel field effect type output powertransistor that is on/off-controlled depending on an output from theSWOUT terminal (23rd pin).

If the transistor Q2 is brought into the on state, a switch currentflows in the coil L1 to the ground terminal via the transistor Q2 andits electric energy is stored. Here, in a case where electric chargesare already accumulated in the output capacitor CVOUT in an on time ofthe transistor Q2, a current from the output capacitor CVOUT flows intoa train of light emitting diodes that are loads (a train of LEDsconnected between a lead-out terminal of the output voltage VOUT and theLED1 terminal to LED4 terminal, which is not shown in FIG. 4, though).Besides, here, because an anode potential of the diode D1 drops toalmost the ground potential via the transistor Q2, the diode D1 goesinto a backward bias state and a current does not flow from the outputcapacitor CVOUT to the transistor Q2.

On the other hand, if the transistor Q2 is brought into the off state,the electric energy accumulated there is discharged by a backwardvoltage generated in the coil L1. Here, because the diode D1 goes to aforward bias state, the current flowing via the diode D1 flows into theLED train that is the load and into the ground terminal as well via theoutput capacitor CVOUT, thereby charging the output capacitor CVOUT. Theabove operation is repeated, so that the LED train, that is, the load,is stepped up in voltage by the output capacitor CVOUT and the smoothedoutput voltage VOUT is supplied.

As described above, the semiconductor device 10 according to the presentembodiment functions as a constituent component of a chopper-typevoltage step-up circuit that drives the coil L1 that is an energystorage element by the on/off control of the transistor Q2, therebystepping up the power-supply voltage VCC to generate the output voltageVOUT.

Next, output feedback control of the voltage step-up DC/DC controllerblock is described in detail.

The error amplifier 113 amplifies a difference between the smallestvalue of the LED terminal voltages V1 to V4 applied respectively to thefirst to fourth inverting input terminals (−) and the predetermined LEDcontrol voltage VLED input into the non-inverting input terminal (+),thereby generating an error voltage Verr. In other words, the voltagevalue of the error voltage Verr goes to a higher level as the outputvoltage VOUT becomes lower than its target set value.

On the other hand, the PWM comparator 108 compares the lower of theerror voltage Verr and an upper limit voltage Vlmt respectively appliedto the first and second non-inverting input terminals (+) with atriangular-wave voltage (lamp-wave voltage) Vosc applied to theinverting input terminal (−), thereby generating a comparison signal(PWM drive waveform) having a duty depending on the comparison result.In other words, the logic of the comparison signal goes to the highlevel if the error voltage Verr (or the upper limit voltage Vlmt) ishigher than the triangular-wave voltage Vosc and goes to the low levelif the error voltage Verr (or the upper limit voltage Vlmt) is lowerthan the triangular-wave voltage Vosc.

Accordingly, the on duty (the ratio of an on time of the transistor Q2per unit time) of the comparison signal in a steady operation timechanges depending on a relative height difference between the errorvoltage Verr and the triangular-wave voltage Vosc.

During a time the above comparison signal is maintained at the highlevel, the control logic portion 109 holds the terminal voltage (i.e.,the gate voltage of the transistor Q2) of the SWOUT terminal at the highlevel via the driver 110. Accordingly, the transistor Q2 is brought intothe on state. On the other hand, during a time the comparison signal ismaintained at the low level, the terminal voltage of the SWOUT terminalis held at the low level. Accordingly, the transistor Q2 is brought intothe off state.

As described above, the voltage step-up DC/DC controller block is sostructured as to perform the drive control of the transistor Q2 based onmonitoring results of the LED terminal voltages V1 to V4 (and the outputvoltage VOUT). Accordingly, it becomes possible to maintain the outputvoltage VOUT at a predetermined value.

Next, the series number of the LED train that is the load is described.

As described above, the voltage step-up DC/DC controller block of thesemiconductor device 10 detects the cathode voltage (i.e., the LEDterminal voltages V1 to V4) of the LED train and controls the outputvoltage VOUT applied to the anode of the LED train so as to allow thecathode voltage to become the LED control voltage VLED (=0.8 [V](Typ.)).

The above voltage step-up operation is performed only when the PWMsignal is in the high level and the output current ILED is flown intothe LED train. Beside, when a plurality of LED trains are driven, theLED terminal voltage (in other words, the smallest value of the LEDterminal voltages) of the LED train that has the largestforward-direction drop voltage VF of the LED is so controlled as tobecome identical to the LED control voltage VLED. Accordingly, the LEDterminal voltages of the other trains of LED terminal voltages become avoltage that is higher by a difference in the forward-direction dropvoltage VF.

Here, a difference allowable voltage Vper (=3.7 [V] (Typ.)) of theforward-direction drop voltage VF is set by the following formula (2)based on a short detection voltage VDSHT (=4.5 [V] (Typ.)) and the LEDcontrol voltage VLED (=0.8 [V] (Typ.)).

Vper=VDSHT−VLED  (2)

Besides, in detecting an open by the open/short detection portion 118,85% of an overvoltage detection reference voltage VDOVP (=2.0 [V](Typ.)) is set as a trigger voltage (open detection voltage VDOP2 (=1.7[V] (Typ.)) (see FIGS. 7 and 10). When this is converted into the outputvoltage VOUT, the maximum value of the output voltage VOUT in the usualoperation time becomes 30.6 [V]=36 [V]×0.85. Accordingly, the seriesnumber N of the LED is limited so as to make the series number N smallerthan a value that is obtained by dividing the maximum value 30.6 [V] ofthe output voltage VOUT by the forward-direction drop voltage VF of oneLED.

Next, the OVP portion 105 is described. A divided voltage VP obtained byresistance-dividing the output voltage VOUT is input into the OVPterminal (25th pin). As described above, based on the series number N ofthe LED train and on the difference allowable voltage Vper of theforward-direction drop voltage VF, it is sufficient to suitably decideon the overvoltage detection reference voltage VDOVP of the OVP portion105 that is compared with this. Besides, in deciding on the overvoltagedetection reference voltage VDOVP, it should be decided on consideringthe open detection voltage VDOP2 (=VDOVP×0.85) as well. Here, after theOVP portion 105 starts once the protection operation, the protectionoperation is released when the output voltage VOUT decreases to 77.5% ofthe overvoltage detection reference voltage VDOVP.

For example, in a case where the resistance values of the resistancedivision circuit are ROVP1 (on the voltage step-up side), ROVP2 (on theGND side), when the output VOUT meets the following formula (3), theprotection operation of the OVP portion 105 starts.

$\begin{matrix}{{VOUT} \geq {\frac{\left( {{{ROVP}\; 1} + {{ROVP}\; 2}} \right)}{{ROVP}\; 2} \times {VDOVP}}} & (3)\end{matrix}$

Here, when ROVP1=330 [kΩ], ROVP2=22 [kΩ], and VDOVP=2.0 [V], theprotection operation of the OVP portion 105 starts at VOUT=32 [V] orhigher

Next, the oscillation frequency FOSC of the voltage step-up DC/DCconverter is described. By externally connecting a pull-down resistor RTwith the RT terminal (26th pin), charge and discharge currents for aninternal capacitor of the oscillator portion 107 are decided on and itis possible to set the oscillation frequency FOSC of the triangular-wavevoltage Vosc. The resistance value of the pull-down resistor RTexternally connected with the RT terminal may be set in view of thefollowing formula (4) and FIG. 15, and a range of 62.6 to 523 [KΩ] isdesirable.

$\begin{matrix}{{{FOSC}\lbrack{kHz}\rbrack} = {\frac{30 \times 10^{6}}{{RT}\lbrack\Omega\rbrack} \times \alpha}} & (4)\end{matrix}$

Here, in the above formula (4), 30×10⁸ [V/A/S] is a constant (±16.6%)that is decided on inside the circuit, and α is a correction coefficient(RT: α=50 [KΩ]: 0.98, 60 [kΩ]: 0.985, 70 [kΩ]: 0.99, 80 [kΩ]: 0.994, 90[kΩ]: 0.996, 100 [kΩ]: 1.0, 150 [kΩ]: 1.01, 200 [kΩ]: 1.02, 300 [kΩ]:1.03, 400 [k Ω]: 1.04, 500 [kΩ]: 1.045).

Besides, in the setting outside the frequency range in FIG. 15, it isnecessary to take care because the switching is likely to stop.

Next, an external synchronization oscillation frequency FSYNC isdescribed. When a clock for external synchronization is input into theSYNC terminal (6th pin) of the voltage step-up DC/DC converter, it isbetter not to perform operations such as switching to internaloscillation and the like during the manipulation. After the input logicof the SYNC terminal is switched from the high level to the low level,it takes a delay time of about 30 [μsec] (Typ.) until an internaloscillation circuit starts to operate. The clock input into the SYNCterminal is valid at only the rising edge. Besides, in a case where theexternal input frequency is lower than the internal oscillationfrequency, because after the above delay time, the internal oscillationcircuit starts to operate, such an input should be avoided.

As described above, in the semiconductor device 10 according to thepresent embodiment, by using the RT terminal or the SYNC terminal, it ispossible to perform variable control of the oscillation frequency FOSCof the voltage step-up DCDC converter block arbitrarily and with greataccuracy. For example, in a case where the semiconductor device 10according to the present embodiment is used as a backlight control meansfor a car navigation monitor, it is possible to avoid the oscillationfrequency FOSC of the voltage step-up DC/DC converter overlapping thefrequency band of radio noise by suitably setting the externalsynchronization oscillation frequency FSYNC from the SYNC terminalmatching the switching control of a radio reception frequency;accordingly, it becomes possible to perform the backlight control of thecar navigation monitor without deteriorating the reception quality ofradio.

Next, the OCP portion 111 is described with reference to FIG. 16.

FIG. 16 is a diagram for describing a connection relationship amongexternal elements related to operation of the OCP portion 111.

As shown in FIG. 16, a detection resistor RCS is inserted between thesource of the power transistor Q2 (N-channel field effect transistor)for the voltage step-up DC/DC converter and the GND, and the connectionnode is connected with the CS terminal (22nd pin).

Besides, to reduce switching noise (spike noise), a low pass filter (aresistor RLPF and a capacitor CLPF) having a cut-off frequency of 1 to 2[MHz] is inserted between the CS terminal and the detection resistorRCS. Here, if the time constant of the low pass filter LPF is too large,the rising of the CS terminal voltage becomes slow, and the detectionoperation of the OCP portion 111 becomes slow; accordingly, for example,it is appropriate that RLPF=100 [Ω], CLPF=1000 [pF] when the oscillationfrequency FOSC=300 [kHz].

Besides, a detection current IOCP in the OCP portion 111 is decided onby the following formula (5) based on the overcurrent protectionoperation voltage VDCS (a constant voltage applied to the non-invertinginput terminal (+) of the comparator 112) and the detection resistorRCS.

IOCP[A]=VDCS(=0.4[V])/RCS[Ω]  (5)

Besides, because the OCP portion 111 is of the pulse-by-pulse type, theSWOUT terminal is fixed at the low level for one period decided on bythe oscillation frequency FOSC of the voltage step-up DC/DC converter;thereafter, the voltage step-up operation is resumed. Besides, because alarge-current line is formed between the detection resistor RCS and theGND, separate wiring to the GND should be performed in the boarddesigning.

Next, the soft start portion 114 is described. In the semiconductordevice 10 according to the present embodiment, the SS terminal (27thpin) is unused and opened. Besides, the open/short detection function ofthe open/short detection portion 118 is masked until the SS terminalvoltage reaches a clamp voltage of 2.5 [V] (Typ.).

Next, detailed description of selection of external components isperformed.

First, selection of the coil L1 is described in detail with reference toFIGS. 17A and 17B.

FIGS. 17A, 17B are each a diagram for describing selection of the coilL1. Here, in FIG. 17A, a ripple component AIL of the coil current IL isshown, and in FIG. 17B, a circuit that constitutes input and outputstages of the DC/DC converter is shown.

The inductor value of the coil L1 extremely influences the ripplecomponent ΔIL (a difference between the maximum value ILMAX and theminimum value ILMIN of the coil current IL) of the coil current IL.Specifically, as shown by the following formula (6), the larger theinductor value of the coil L1 is, and the higher the oscillationfrequency FOSC is, the smaller the ripple component ΔIL becomes.

$\begin{matrix}{{\Delta \; {{IL}\lbrack A\rbrack}} = \frac{\left( {{VOUT} - {VCC}} \right) \times {VCC}}{L\; 1 \times {VOUT} \times {FOSC}}} & (6)\end{matrix}$

Besides, when the efficiency η is represented as shown by the followingformula (7a), the maximum value ILMAX of the coil current IL becomes asshown by the formula (7b).

$\begin{matrix}{\eta = \frac{{VOUT} \times {IOUT}}{{VCC} \times {ICC}}} & \left( {7a} \right) \\{{{ILMAX}\lbrack A\rbrack} = {{{ICC} + \frac{\Delta \; {IL}}{2}} = {\frac{{VOUT} \times {IOUT}}{{VCC} \times \eta} + \frac{\Delta \; {IL}}{2}}}} & \left( {7b} \right)\end{matrix}$

If the coil current IL exceeding the rated current value of the coil L1is flown in the coil L1, the coil L1 reaches magnetic saturation and theefficiency η decreases. Accordingly, the coil L1 should be selected witha sufficient margin so as not to allow the maximum value ILMAX of thecoil current IL to exceed the rated current value of the coil L1.Besides, to reduce loss in the coil L1 and improve the efficiency η, asthe coil L1, a coil that has a small resistance component (adirect-current reactor DCR, an alternating-current ACR) should beselected.

Next, detailed description of selection of the output capacitor CVOUT isperformed with reference to FIG. 18.

FIG. 18 is a diagram for describing selection of the output capacitorCVOUT, and a circuit that constitutes the input and output stages of theDC/DC converter is shown.

As for the selection of the output capacitor CVOUT, in light of a stabledomain of the output voltage VOUT and further considering an equivalentseries resistance ESR necessary to smooth a ripple component ΔVOUT ofthe output voltage, it is sufficient to suitably decide on the outputcapacitor CVOUT.

The ripple component ΔVOUT of the output voltage VOUT is decided on asshown by the following formula (8).

            (8)${\Delta \; {{VOUT}\lbrack V\rbrack}} = {{{ILMAX} \times {RESR}} + {\frac{1}{CVOUT} \times \frac{IOUT}{\eta} \times \frac{1}{FOSC}}}$

Here, in the above formula (8), ΔIL represents the ripple component ofthe output current IL; RESR represents the resistance value of theequivalent series resistance ESR of the output capacitor CVOUT; and ηrepresents the efficiency.

Here, it is desirable to select the rating of the output capacitor CVOUTwith a sufficient margin with respect to the output voltage VOUT.

Next, selection of the input capacitor CVCC is described in detail withreference to FIG. 19.

FIG. 19 is a diagram for describing selection of the input capacitorCVCC, and a circuit that constitutes the input and output stages of theDC/DC converter is shown.

As for the selection of the input capacitor CVCC, it is desirable to usean input capacitor having a low ESR that has a capacitance value capableof sufficiently interacting with a large ripple current IRMS so as toprevent a large transient voltage.

The above ripple current IRMS is given by the following formula (9).

$\begin{matrix}{{{IRMS}\lbrack A\rbrack} = {{IOUT} \times \frac{\sqrt{\left( {{VOUT} - {VCC}} \right) \times {VOUT}}}{VOUT}}} & (9)\end{matrix}$

Besides, because the ripple current IRMS extremely depends on thecharacteristics of the power supply used for the input, the wiringpattern of the board and the gate-drain capacitances of the transistorsQ1, Q2, it is desirable to sufficiently confirm the temperature, theload range, and the conditions of the transistors Q1, Q2 in a time ofuse.

Next, selection of the load-switch transistor Q1 and its soft start aredescribed in detail with reference to FIGS. 20 and 21.

FIG. 20 is a diagram for describing selection of the load-switchtransistor Q1 and its soft start, and a circuit that constitutes theinput and output stages of the DC/DC converter is shown. Besides, FIG.21 is a graph showing a correlation between the gate-source capacitanceof the transistor Q1 and the soft start time.

In a case of a usual voltage step-up application, there is no switch onthe route that extends from the application terminal of the power-supplyvoltage VCC to the lead-out terminal of the output voltage VOUT;accordingly, if an output short-circuit occurs, an overcurrent flows inthe route, so that the coil L1 and the rectifying diode D1 can bebroken. To avoid this, in the semiconductor device 10 according to thepresent embodiment, the P-channel type field effect transistor Q1 forthe load-switch is inserted between the application terminal of thepower-supply voltage VCC and the coil L1. Here, as the transistor Q1, itis sufficient to select a transistor whose gate-source breakdown voltageand whose drain-source breakdown voltage are both higher than thepower-supply voltage VCC.

Besides, to perform the soft start of the load switch, it is sufficientto insert capacitance between the gate and source of the transistor Q1.According to this, as shown in FIG. 21, it is possible arbitrarily todecide on the soft start time depending on the inserted capacitancevalue. However, the soft start time changes depending on the gatecapacitance of the transistor Q1 as well.

Next, selection of the switching transistor Q2 is described. There is noproblem whatever MOSFET is used if the MOSFET has the absolute maximumrated current that is larger than the rated current of the coil L1 andthe absolute maximum rated voltage that is higher than the breakdownvoltage of the output capacitor CVOUT+the forward-direction drop voltageVF of the rectifying diode D1; however, to achieve high-speed switching,a MOSFET that has a small gate capacitance (the amount of injectedelectric charges) should be selected, and it is desirable to use aMOSFET that has gate capacitance lager than the overcurrent protectionset value. Besides, if a MOSFET that has a small on resistance isselected, it becomes possible to obtain high efficiency.

Next, selection of the rectifying diode D1 is described. Whatever diodemay be used if the diode is a Schottky barrier diode that has anelectric-current capability equal to or larger than the rated current ofthe coil L1 and a backward breakdown voltage equal to or higher than thebreakdown voltage of the output capacitor CVOUT; especially, it issufficient select a diode whose forward-direction drop voltage VF islow.

Next, detailed description of a phase-compensation setting method isperformed.

First, stable conditions of an application are described. As forconditions under which a feedback system with negative feedback returnedbecomes stable, it is necessary that the phase delay is 150° or smaller(i.e., the phase margin is 30° or larger) at a time the gain is 1(0[dB]).

Besides, because the DC/DC converter application is sampled by theoscillation frequency FOSC, it is necessary to set the GBW (Gain-BandWidth) (a product of the gain and the band width) of the entire systemat a value that is 1/10 the oscillation frequency FOSC or lower.

Summing up the above description, the characteristics the applicationaims at are the phase delay that is 150° or smaller (i.e., the phasemargin is 30° or larger) at the time the gain is 1 (0 [dB]); and it issufficient if the GBW (i.e., the frequency at the gain of 0 [dB]) atthat time is 1/10 the oscillation frequency FOSC or lower. Accordingly,to improve the response according to a limit to the GBW, it is necessaryto raise the oscillation frequency FOSC.

To secure the stability by phase compensation, it is sufficient tocancel a secondary phase delay (−180°) due to an LC resonance by asecondary phase advance (i.e., two phase advances are used). Here, as ameans for giving a phase advance, the ESR component (see FIG. 23) of theoutput capacitor CVOUT and the CR component (see FIG. 24) connected withthe COMP terminal (28th pin) are possible.

In the DC/DC converter application, as shown in FIG. 22, there isinvariably an LC resonance circuit. Accordingly, the phase delay at thatportion becomes −180°. As shown in FIG. 23, in a case where the outputcapacitor CVOUT is an element such as an aluminum electrolytic capacitoror the like that has a large ESR (a few ohms [Ω]), a phase advance of+90° occurs and the phase delay becomes −90°. On the other hand, in acase where the output capacitor CVOUT such as a ceramic capacitor or thelike that has a low ESR is used, it is necessary to insert resistanceequal to the ESR component.

Here, because of a change in the phase characteristic due to the ESR,the phase advance to be inserted becomes one. Besides, as for setting ofthe frequency into which a phase advance is inserted, for the purpose ofcanceling the LC resonance, it is ideally desirable to set the frequencynear the LC resonance frequency.

Next, an operation sequence of the semiconductor device 10 is describedwith reference to FIG. 25.

FIG. 25 is a timing chart showing an operation sequence of thesemiconductor device 10.

When the enable signal EN rises to high level after the power-supplyvoltage VCC is turned on, generation of the reference voltage VREG isstarted in the VREG portion 101. Here, as for the enable signal EN,after the power-supply voltage VCC sufficiently rises, for example,after the power-supply voltage VCC becomes 4.5 [V] or higher, it issufficient to turn on the enable signal EN.

When the reference voltage VREG reaches 2.9 [V], it is recognized by theUVLO portion 103 that it is not a reduced-voltage state, and the UVLOsignal rises to the high level. According to this, the internal circuitof the semiconductor device 10 starts to operate. Here, during a timethe UVLO signal is maintained in the low level, the switch 102 ismaintained in the off state and the terminal voltage of the LOADSWterminal (2nd pin) is maintained in the high level. Accordingly, becausethe load-switch transistor Q1 is turned off, the voltage step-upoperation of the DC/DC converter is maintained in a stop state. On theother hand, when the UVLO signal rises to the high level, the switch 102is turned on, and the terminal voltage of the LOADSW terminal falls tothe low level. As a result of this, the load-switch transistor Q1 isturned on, and the voltage step-up operation becomes possible.

For stable operation, there are predetermined input sequences for theVDAC signal, SYNC signal and PWM signal that are external signals.Specifically, it is desirable to input the VDAC signal and SYNC signalafter a first predetermined time TINON elapses from the input timing ofthe enable signal EN; and it is desirable to input the PWM signal aftera second predetermined time TPWMON elapses from the input timing of theEN signal. Here, the second predetermined time TPWMON>the firstpretermined time TINON and the second predetermined time TPWMON>500[V/A·s]×CREG [sec]. Besides, it is desirable to block the inputs of theVDAC signal and SYNC signal earlier than the EN signal by a thirdpredetermined time TINOFF, while it is desirable to block the input ofthe PWM signal earlier than the EN signal by a fourth predetermined timeTPWMOFF. Here, the fourth predetermined time TPWMOFF>the thirdpredetermined time TINOFF. Besides, although not shown in this figure,it is desirable fix the logics of the LEDEN1 signal and LEDEN2 signalbefore the EN signal shifts to the high level.

In the OVP portion 105, when the terminal voltage of the OVP terminal(25th pin) reaches 2 [V], it is recognized as an overvoltage state, andthe voltage step-up operation is stopped. Thereafter, in the OVP portion105, when the terminal voltage of the OVP terminal drops to 1.6 [V], itis recognized that the overvoltage state is released, and the voltagestep-up operation is resumed.

In the OCP portion 111, when the terminal voltage of the CS terminal(22nd pin) reaches 0.4 [V], it is recognized as an overcurrent state,and thereafter, the voltage step-up operation of the DC/DC converter isintermittently turned on/off by the pulse-by-pulse fashion.

In the TSD portion 104, when the temperature of the semiconductor device10 reaches 175[° C.], it is recognized as an abnormal heating state, andthe voltage step-up operation of the DC/DC converter is stopped.Thereafter, in the TSD portion 104, when the temperature of thesemiconductor device 10 drops to 150[° C.], it is recognized that theabnormal heating state is released, and the voltage step-up operation isresumed.

Here, if the EN signal is made fall to the low level, the generation ofthe reference voltage VREG is stopped. In the UVLO portion 103, whenthis reference voltage VREG drops to 2.8 [V], it is recognized as areduced-voltage state, and the UVLO signal falls to the low level. Inthis way, the internal circuit of the semiconductor device 10 stops theoperation.

Next, detailed description of a second embodiment of the semiconductordevice according to the present invention is performed.

FIG. 26 is a block diagram showing a second embodiment of thesemiconductor device according to the present invention.

First, an overview of a semiconductor device 20 according to the presentembodiment is described.

The semiconductor device 20 is a 36 V-resistant white-LED driver IC; anda voltage step-up and down DC/DC converter of a current mode and afour-channel output constant-current driver are integrated into onechip. The semiconductor device 20 is able to perform light control ofthe white LED by using any of PWM [Pulse Width Modulation] control andVDAC control.

Next, of features of the semiconductor device 20 according to thepresent embodiment, especially, points different from the firstembodiment is described.

A first feature is that to interact with the power-supply voltage VCCdirectly supplied from a battery, a voltage step-up/-down DC/DCcontroller block is incorporated instead of the voltage step-up DC/DCcontroller block. A second feature is that to use a low-ESR ceramiccapacitor as the output capacitor CVOUT, the control mo de of the DC/DCconverter is changed from a voltage mode to an electric-current mode. Athird feature is that to raise the PWM light control capability of LEDluminous brightness, a duty ratio (with no overshoot) of 0.38[%] isachieved. A fourth feature is that a relative uneven width ±3% of theoutput current ILED and an absolute uneven width ±5% of the outputcurrent ILED are achieved. A fifth feature is that a protection functionportion (SCP [Short Circuit Protection]) which detects a short in theanode and cathode of the LED and performs an appropriate protectionoperation is incorporated.

Here, the semiconductor device 20 according to the present embodiment isused for drive control of a backlight of a car navigation monitor,backlights of medium- and small-sized LED panels and the like.

The semiconductor device 20 according to the present embodiment havingthe above features, as shown in FIG. 26, is composed of an integrationof: a reference voltage generation portion 201 (hereinafter, called aVREG portion 201); a reduced-voltage protection portion 202(hereinafter, called a UVLO portion 202); a temperature protectionportion 203 (hereinafter, called a TSD portion 203); a short protectionportion 204 (hereinafter, called a SCP portion 204); an overvoltageprotection portion 205 (hereinafter, called an OVP portion 205); anovercurrent protection portion 206 (hereinafter, called an OCP portion206); a comparator 207; a control logic portion 208; an input buffer209; an oscillator portion 210; a slope-voltage generation portion 211;a PWM comparator 212; a driver control portion 213; a driver 214; anN-channel type field effect transistor 215; a driver 216; an erroramplifier 217; a soft start portion 218; an input buffer 219; a currentset portion 220; a constant-current driver 221; an open/short detectionportion 222; and input buffers 223 and 224.

Here, it is possible to roughly divide the above circuit portion of thesemiconductor device 20 into four blocks of: a VREG block (the VREGportion 201); a voltage step-up and -down DC/DC controller block (theOCP portion 206, comparator 207, control logic portion 208, input buffer209, oscillator portion 210, slope-voltage generation portion 211, PWMcomparator 212, driver control portion 213, driver 214, transistor 215,driver 216, error amplifier 217 and soft start portion 218); a currentdriver block (the input buffer 219, current set portion 220,constant-current driver 221, open/short detection portion 222 and inputbuffers 223 and 224); and a protection block (the UVLO portion 202, TSDportion 203, SCP portion 204 and OVP portion 205).

Besides, the semiconductor device 20 according to the present embodimentincludes 28 external terminals (1st to 28th pins) as means for securingelectric connections with outside.

FIG. 27 is a pin arrangement diagram of the semiconductor device 20 andFIG. 28 is a table which shows pin numbers, terminal names and functionsof the external terminals. In FIG. 27, wide terminals disposed at bothsides of central portions of the semiconductor device 20 are FINterminals that are linked to subground and improve heat radiation.

FIG. 29 is a table showing electric characteristics of the semiconductordevice 20 that has the above structure. Here, the electriccharacteristics shown in FIG. 29 represent numerical values at thepower-supply voltage VCC=12 [V], ambient temperature Ta=25[° C.] unlessotherwise specified.

Next, detailed description of operation of each portion of thesemiconductor device 20 is performed centering on points different fromthe first embodiment.

First, detailed description of the current driver block of thesemiconductor device 20 (the input buffer 219, current set portion 220,constant-current driver 221, open/short detection portion 222 and inputbuffers 223 and 224) is performed.

FIG. 30 is a circuit diagram showing structural examples of the currentset portion 220 and the constant-current driver 221.

As shown in FIG. 30, the current set portion 220 includes: theoperational amplifier A1; the direct-current voltage source A2; thenpn-type bipolar transistor A3; the resistors A4, A5; the pnp-typebipolar transistors A6 to A9; and the resistor A10 (the resistance valueR).

The first non-inverting input terminal (+) of the operational amplifierA1 is connected with the VDAC terminal (8th pin). The secondnon-inverting input terminal (+) of the operational amplifier A2 isconnected with the positive-polar terminal of the direct-current voltagesource A2, and the predetermined constant voltage VISET (=2.0 [V]) isapplied. The negative-polar terminal of the direct-current voltagesource A2 is connected with the ground terminal. The inverting inputterminal (−) of the operational amplifier A2 is connected with the ISETterminal (9th pin). The base of the transistor A3 is connected with theoutput terminal of the operational amplifier A1. The emitter of thetransistor A3 is connected with the ISET terminal.

One terminal of each of the resistors A4, A5 is connected with theapplication terminal of the reference voltage VREG. The other terminalof the resistor A4 is connected with the emitter of the transistor A6.The other terminal of the resistor A5 is connected with the emitter ofthe transistor A7. The bases of the transistors A6, A7 are connectedwith each other and the connection node is connected with the collectorof the transistor A7. The collector of the transistor A6 is connectedwith the emitter of the transistor A8. The collector of the transistorA7 is connected with the emitter of the transistor A9. The bases of thetransistors A8, A9 are connected with each other and the connection nodeis connected with the collector of the transistor A8. The collector ofthe transistor A8 is connected with the collector of the transistor A3.The collector of the transistor A9 is connected with the ground terminalvia the resistor A10.

On the other hand, as shown in FIG. 30, the constant-current driver 221includes 4 channels of output stages Ch1, Ch2, Ch3 and Ch4 that supplythe output current ILED to the LED1 terminal to LED4 terminal,respectively. Here, the output stage Ch1 includes: the operationalamplifier B1; the N-channel type field effect transistor B2; theresistor B3 (the resistance value 4R); the current mirror circuit B4(the mirror ratio 1:1); the resistor B5 (the resistance value 4R); theoperational amplifier B6; the N-channel type field effect transistor B7;the resistor B8 (the resistance value ( 4/12)×R); the current mirrorcircuit B9 (the mirror ratio 1:10); the operational amplifier B10; thedirect-current voltage source B11; the N-channel type field effecttransistors B12 to B14; the operational amplifier B15; thedirect-current voltage source B16; the N-channel type field effecttransistor B17; an N-channel type field effect transistor B19; aP-channel type field effect transistor B20; resistors B21, B22; anN-channel type field effect transistor B23; and an inverter B24.

The non-inverting input terminal (+) of the operational amplifier B1 isconnected the connection node of the transistor A9 and the resistor A10.The inverting input terminal (−) of the operational amplifier B1 isconnected with one terminal of the resistor B3. The other terminal ofthe resistor B3 is connected with the ground terminal. The drain of thetransistor B2 is connected with the input terminal of the current mirrorcircuit B4. The source of the transistor B2 is connected with oneterminal of the resistor B3. The gate of the transistor B2 is connectedwith the output terminal of the operational amplifier B1. Thepower-supply input terminal of the current mirror circuit B4 isconnected with the application terminal of the reference voltage VREG.

The non-inverting input terminal (+) of the operational amplifier B6 isconnected with the output terminal of the current mirror circuit B4 andwith one terminal of the resistor B5. The inverting input terminal ofthe operational amplifier B6 is connected with one terminal of theresistor B8. Both of the other terminals of the resistors B5, B8 areconnected with the ground terminal. The drain of the transistor B7 isconnected with the input terminal of the current mirror circuit B9. Thesource of the transistor B7 is connected with one terminal of theresistor B8. The gate of the transistor B7 is connected with the outputterminal of the operational amplifier B6. The power-supply inputterminal of the current mirror circuit B9 is connected with theapplication terminal of the reference voltage VREG.

The non-inverting input terminal (+) of the operational amplifier B10 isconnected with the positive-polar terminal of the direct-current voltagesource B11. The negative-polar terminal of the direct-current voltagesource B11 is connected with the ground terminal. The drain of thetransistor B12 is connected with the output terminal of the currentmirror circuit B9. The source of the transistor B12 is connected withthe inverting input terminal (−) of the operational amplifier B10. Thegate of the transistor B12 is connected with the output terminal of theoperational amplifier B10.

The drain of the transistor B13 is connected with the source of thetransistor B12. The gates of the transistors B13, B14 are connected witheach other and the connection node is connected with the drain of thetransistor B12 and also connected with the drain of the transistor B19.All the sources of the transistors B13, B14 and B19 are connected withthe ground terminal. The gate of the transistor B19 is connected withthe PWM terminal (8th pin) via the input buffer 219 (not shown in thisfigure).

The non-inverting input terminal (+) of the operational amplifier B15 isconnected with the positive-polar terminal of the direct-current voltagesource B16. The negative-polar terminal of the direct-current voltagesource B16 is connected with the ground terminal. The drain of thetransistor B17 is connected with the LED1 terminal. The source of thetransistor B12 is connected with the inverting input terminal (−) of theoperational amplifier B15 and also connected with the drain of thetransistor B14. The gate of the transistor B17 is connected with theoutput terminal of the operational amplifier B15.

A source of the transistor B20 is connected with the applicationterminal of the reference voltage VREG. A drain of the transistor B20 isconnected with the input terminal of the current mirror circuit B9. Oneterminal of the resistor B21 is connected with the application terminalof the reference voltage VREG. The other terminal of the resistor B21 isconnected with a gate of the transistor B20. One terminal of theresistor B22 is connected with the gate of the transistor B20. The otherterminal of the resistor B22 is connected with a drain of the transistorB23. A source of the transistor B23 is connected with the groundterminal. A gate of the transistor B23 is connected with an outputterminal of the inverter B24. An input terminal of the inverter B24 isconnected with the PWM terminal via the inut buffer 219 (not shown inthis figure).

Here, because the other output stages Ch2 to Ch4 that constitute theconstant-current driver 221 include the same structure as that of theabove output stage Ch1, detailed description of them is skipped.

In the current set portion 220 and the constant-current driver 221 thathave the above structures, the output current ILED is set based on thefollowing formula (10).

ILED[mA]=min{VDAC,2.0[V]}/RSET[kΩ]×3000  (10)

In the above formula (10), the parameter min {VDAC, 2.0 [V]} is avoltage value that is the lower of the control voltage VDAC input intothe VDAC terminal (18th pin) and the constant voltage VISET (=2.0 [V])predetermined in the current set portion 220. Besides, the parameterRSET is a resistance value of the resistor RSET that is externallyconnected with the ISET terminal (19th pin); and a parameter 3000 (Typ.)is a constant that is decided on in the constant-current driver 221.

Specifically, the resistor RSET is pulldown-connected with the ISETterminal (19th pin), so that an electric current predetermined-gaintimes (e.g., 3000 times) higher than the reference current ISET flowingin this is set as the maximum value (e.g., 50 [mA]) of the outputcurrent ILED.

Describing specifically with reference to the example in FIG. 30, in theconstant-current driver 221, first, by using the operational amplifierB1, transistor B2 and resistor B3 (the resistance value 4R), theterminal voltage Va (=ISET×R) of the resistor A10 isvoltage/current-converted to generate the intermediate current Ia(=1/4ISET) that is ¼ the reference current ISET. Next, by using thecurrent mirror circuit B4, the intermediate current Ia is mirrored at1:1 to generate the intermediate current Ib (=1/4ISET). Next, by usingthe resistor B5 (the resistance value 4R), the intermediate Ib iscurrent/voltage-converted to generate the terminal voltage Vb (=ISET×R).Next, by using the operational amplifier B6, transistor B7 and resistorB8 (the resistance value ( 4/12)×R)), the terminal voltage Vb of theresistor B5 is voltage/current-converted to generate the intermediatecurrent Ic (=3ISET) that is times as large as the reference currentISET. Next, by using the current mirror circuit B9, the intermediatecurrent Ic is mirrored at 1:10 to generate the intermediate current Id(=30ISET) that is 30 times as large as the reference current ISET. And,finally, by using the current mirror circuit that includes thetransistors B13, B14, the intermediate current Id is mirrored at 1:100to generate the output current ILED (=3000ISET) that is 3000 times aslarge as the reference current ISET.

Here, to raise the accuracy of the output current ILED, in thelast-stage current mirror circuit, by using the operational amplifiersB10, B15, the drain-source voltages of the transistors B13, B14 are madeidentical to each other (e.g., 0.3 [V]). Besides, the constant-currentdriver 221 is so structured as to generate the desired output currentILED by repeating the voltage/current conversion and the current/voltageconversion based on the input reference current ISET. Accordingly,resistor elements (the resistors B3, B5 and B8 in the example in FIG.30) used for the above conversion processes increase and the trimmingchances increase. As described above, according to the structureincluding many resistors that are able to be trimmed, by finelyadjusting the resistance values, it becomes possible to reduce therelative uneven width and the absolute uneven width of the outputcurrent ILED.

Unlike the first embodiment that amplitudes the reference current ISETwithout stopping by the last-stage current mirror circuit(ISET→1/5ISET→2ISET→3300ISET), the semiconductor device 20 according tothe second embodiment is so structured as to dispersedly perform currentamplification in the generation process of the output current ILED inthe constant-current driver 221 (ISET1/4ISET3ISET→30ISET→3000ISET).

According to such structure, as shown by a comparison of FIG. 31A andFIG. 31B, it is possible to reduce the unevenness in the production ofelements and the influence of stress by decreasing a difference betweenthe transistor sizes of the transistors B13, B14 that constitute thelast-stage current mirror circuit. Specifically, if the differencebetween the transistor sizes is large, a state in which a large stressacts on only one transistor while almost no stress acts on the othertransistor occurs, that is, the transistors are likely to be subjectedto the influence of stress that is generated during a packaging time ofthe semiconductor device 20; however, according to the structure in thepresent embodiment, because the difference between the transistor sizesbecomes small, an equal stress is likely to act on both elements, sothat it becomes possible to reduce the influence of stress. Of course,in designing the transistor elements, it is desirable to suitably designW/L of each element to allow the transistors to operate in a saturationdomain in an actual use range (5 [mA] or more is expected) of the outputcurrent ILED.

Besides, in the semiconductor device 20 according to the presentembodiment, as shown in FIG. 32, pairs of resistors R1, R2 (e.g., theresistors B3 and B5 or the resistors B5 and B8) are arranged in a zigzaglayout. According to the employment of such arrangement layout, inpackaging the semiconductor device 20, an equal stress is likely to acton the pair of resistors R1, R2, so that it becomes possible to reducethe influence of stress.

As described above, in the semiconductor device 20 according to thepresent embodiment, the accuracy of the output current ILED is improvedfrom many sides, that is, the increase in the resistor trimming chance,dispersion of the current amplification capability, improvement in thepairing easiness of resistors and the like. According to such structure,it is possible to achieve the relative uneven width of ±3% and theabsolute uneven width of ±5% of the output current ILED after thepackaging and it becomes possible to contribute to reduction in thebrightness unevenness and longevity of the LED.

Here, as for the resistance value of the resistor RSET, as describedwith reference to the above FIG. 11, it is desirable to use a resistorthat has a resistance of 300 [kΩ] or lower.

Here, in a case where the variable control (light control of the LED) ofthe output current ILED is performed by using the above control voltageVDAC, it is sufficient to set the input range at a range of 0.1 to 2.0[V]. By applying such control voltage VDAC, it becomes possible todecrease the output current ILED from the maximum value.

On the other hand, in a case where 2.0 [V] or higher is input as thecontrol voltage VDAC, as given by the above formula (10), the voltagevalue of the constant voltage VISET is selected; accordingly, a non-usestate in which the light control function by the control voltage VDAC isnot used occurs. In a case where the light control by the controlvoltage VDAC is not used, from the viewpoint of avoidance ofmalfunction, it is sufficient not to open the VDAC terminal but connectit with the application terminal of the reference voltage VREG (5 [V]).

In addition, in the semiconductor device 20 according the presentembodiment, besides the light control of the LED that uses the abovecontrol voltage VDAC, by using the PWM signal input into the PWMterminal (8th pin), the on/off (in the example in FIG. 30, the on/off ofthe transistors B13, B14 that constitute the last-stage current mirrorcircuit and the on/off of the current mirror circuit B9) of theconstant-current driver 221 is controlled, so that it is possible toperform the light control of the LED as well.

Specifically, in the semiconductor device 20 according to the presentembodiment, the on/off of the transistors B13, B14 that constitute thelast-stage current mirror circuit and the on/off of the current mirrorcircuit B9 are controlled based on the PWM signal, so that the dutyratio of the PWM signal becomes the duty ratio of the output currentILED; accordingly, it becomes possible to seemingly decrease the outputcurrent ILED from the maximum value (or a current value decided on bythe control voltage VDAC).

Here, in the first embodiment that converts the reference current ISETinto a pulse current based on the PWM signal, ringing occurs in thecurrent mirror circuits B4, B9, which leads to an overshoot of theoutput current ILED; however, in the structure according to the presentembodiment, such problem does not occur.

Besides, in the semiconductor 20 according to the present embodiment, asa further measure against an overshoot, an operational amplifier thathas a slow slew rate (e.g., 0.5 [V/μs]) is used as the output-stageoperational amplifier B15; and fluctuation in the gate-source voltageVGS of the transistor B17 is limited, so that the rising of the outputcurrent ILED is slowed down to prevent an overshoot from occurring.

Further, in the semiconductor device 20 according to the presentembodiment, as described above, because the intermediate current Id thatflows into the last-stage current mirror circuit is increased from 2ISETin the first embodiment to 30ISET, the response of the transistors B13,B14 rises, which allows removal of the pull-down resistor B18;accordingly, it becomes possible to further curb occurrence of anovershoot.

As described above, in the semiconductor device 20 according to thepresent embodiment, from many sides such as the PWM control of thelast-stage current mirror circuit, use of the operational amplifier B15having a slow slew rate, removal of the pull-down resistor B18 and thelike, the improvement in response of the output current TIED and thereduction in overshoot are performed. According to such structure,without causing an overshoot, it becomes possible to achieve improvement(the minimum duty ratio: 0.38% (at 150 [Hz])) in the PWM light controlcapability and improve the light control accuracy at a low duty.

On the other hand, in a case where the light control function by the PWMsignal is not used, it is sufficient to fix the PWM terminal at the highlevel (e.g., the constant voltage VREG). Here, it is desirable to inseta low pass filter (a cut-off frequency of 30 [kHz]) into the PWMterminal.

An example of the PWM light control is already described with referenceto the above FIGS. 14A to 14C.

Next, the voltage step-up and down DC/DC controller block of thesemiconductor device 20 (the OCP portion 206, comparator 207, controllogic portion 208, input buffer 209, oscillator portion 210,slope-voltage generation portion 211, PWM comparator 212, driver controlportion 213, driver 214, transistor 215, driver 216, error amplifier 217and soft start portion 218) is described in detail with reference to theabove FIG. 26.

First, detailed description of external connections of the semiconductordevice 20, especially circuit elements (N-channel type field effecttransistors N1, N2, diodes D2, D3, coil L2, resistors RCS, RLPF,capacitors CBS, CLPF) related to the voltage step-up and down DC/DCconverter is performed.

As shown in FIG. 26, a gate of the transistor N1 is connected with anoutput terminal of the driver 214 via the OUTH terminal (25th pin). Adrain of the transistor N1 is connected with the application terminal ofthe power-supply voltage VCC via the resistor RCS and also connectedwith the CS terminal (27th pin) via the resistor RLPF. A source of thetransistor N1 is connected with a second power-supply terminal(low-potential terminal) of the driver 214 and a drain of the transistor215 via the SW terminal (24th pin).

One terminal of the coil L2 is connected with the SW terminal. The otherterminal of the coil L2 is connected with an anode of the diode D3. As alead-out terminal of the output voltage VOUT, a cathode of the diode D3is connected with the anode of the LED train that is the load.

A gate of the transistor N1 is connected with an output terminal of thedriver 216 via the OUTL terminal (22nd pin). A drain of the transistorN2 is connected with a connection node of the other terminal of the coilL2 and the anode of the diode D3. A source of the transistor N3 isconnected with the ground terminal.

A cathode of the diode D2 is connected with SW terminal. An anode of thediode D2 is connected with the ground terminal. One terminal of thecapacitor CBS for bootstrap is connected with a first power-supplyterminal (high-potential terminal) of the driver 214 via the BOOTterminal (9th pin). The other terminal of the capacitor CBS is connectedwith the SW terminal. One terminal of the capacitor CLPF is connectedwith the application terminal of the power-supply voltage VCC. The otherterminal of the capacitor CLPF is connected with the CS terminal.

Here, in the semiconductor device 20 according to the presentembodiment, because the transistors N1, N2 are externally connected, itbecomes possible to raise heat radiation.

Next, detailed description of basic operation of the voltage step-up and-down DC/DC controller block is performed.

If the transistors N1, N2 are brought into an on state, a current flowsinto the coil L2 via a route X and its electric energy is stored.Besides, in a case where electric charges are accumulated in thecapacitor CVOUT in an on time of the transistors N1, N2, a current fromthe capacitor CVOUT flows into the lead-out terminal of the outputvoltage VOUT. Here, because the other-terminal potential of the coil L2drops to almost the ground potential via the transistor N2, the diode D3goes into a backward bias state and a current does not flow from thecapacitor CVOUT to the transistor N2.

Next, if the transistors N1, N2 are brought into an off state, theelectric energy accumulated there is discharged by a backward voltagegenerated in the coil L1 via a route Y, flows from the lead-out terminalof the output voltage VOUT into the LED train that is the load, and alsoflows into the ground terminal via the capacitor CVOUT to charge thecapacitor CVOUT.

Accordingly, in the semiconductor device 20 according to the presentembodiment, by suitably controlling the duty ratios of the transistorsN1, N2 by means of the driver control portion 213, specifically, in atime of voltage step-down operation, the duty ratios of the transistorsN1, N2 are decreased to a value smaller than 50%, and to the contrary,in a time of voltage step-up operation, the duty ratios of thetransistors N1, N2 are increased to a value larger than 50%, so that itbecomes possible to easily and suitably change the voltage step-up and-down operations even with a simple structure.

In other words, in the semiconductor device 20 according to the presentembodiment, regardless of whether the power-supply voltage VCC is ishigher or lower than the desired output voltage VOUT, it becomespossible to always obtain the desired output voltage VOUT. Accordingly,for example, even in a case where the power-supply voltage VCC changesin a range of 6 to 18 [V] when the desired value of the output voltageis 16 [V], it becomes possible to obtain the desired output voltageVOUT. Such structure is suitable for an application, for example, (e.g.,a backlight-control LED driver IC for a car navigation monitor) thatneeds to interact with the power-supply voltage VCC which is directlysupplied from a battery.

Note that even in a structure in which the gate of the transistor N2 isconnected with the SW terminal, the above voltage step-un and -downoperations are possible; however, the semiconductor device 20 accordingto the present embodiment achieves the breakdown voltage of 36 V, andsuch a high voltage is likely to be applied to the SW terminal as well.On the other hand, the gate breakdown voltage of the external transistorN2 is not invariably high. Accordingly, in a structure in which the gateof the transistor N2 is connected with the SW terminal, the transistorN2 can be broken.

Because of this, in the semiconductor device 20 according to the presentembodiment, as a means to perform the gate control of the transistor N2,the separate driver 216 (which operates on the reference voltage VREG)is prepared, and by using this, the on/off control of the transistor isperformed. According to such structure, there is no worry over that thetransistor N2 is broken even if the power-supply voltage is a highvoltage.

Besides, in the semiconductor device 20 according to the presentembodiment, as a ringing protection means in a time of a light load orno load, the N-channel type field effect transistor 215 is integrated.

A drain of the transistor 215 is connected with the SW terminal. Asource of the transistor 215 is connected with the ground terminal. Agate of the transistor 215 is connected with a control signal outputterminal of the driver control portion 213.

Here, it is desirable that not to cause increase in an unnecessary chiparea nor decrease in conversion efficiency, the electric-currentcapability of the transistor 215 is designed to be the smallest possiblecapability of removing a small current of ringing noise.

The driver control portion 213 performs the on/off control of thetransistors N1, N2, while it performs the on/off control of thetransistor 215 in a complementary fashion to this.

According to such structure, even in a case where the output currentdrops and the coil current is entirely lowered in a light-load orno-load time, and is trapped into a state (so-called discontinuous mode)in which ringing, that is, deformation in a waveform occurs, it ispossible to enable the ringing noise to escape to the ground terminalvia the transistor 215, so that it becomes possible to raise thestability of the voltage step-up and -down operations.

Here, the language “complementary” used in the above description coversnot only a case where the on/off of the transistor N1 (and thetransistor N2) and the on/off of the transistor 215 are completelyopposite to each other but also a case where from viewpoints ofprevention of a through current and the like, a predetermined delay isgiven to the on/off transition timing of the transistor N1 and thetransistor 215.

Next, output feedback control of the voltage step-up DC/DC controllerblock is described in detail.

The error amplifier 217 amplifies a difference between the smallestvalue of the LED terminal voltages V1 to V4 applied respectively to thefirst to fourth inverting input terminals (−) and the predetermined LEDcontrol voltage VLED input into the non-inverting input terminal (+),thereby generating an error voltage Verr. In other words, the voltagevalue of the error voltage Verr goes to a higher level as the outputvoltage VOUT becomes lower than its target set value.

On the other hand, the PWM comparator 212 compares the error voltageVerr applied to the non-inverting input terminal (+) with a slopevoltage Vslp (a sum voltage of the triangular-wave voltage Vosc of theoscillator OSC and the terminal voltage (an electric-current detectionsignal generated at the resistor RCS) of the CS terminal (27th pin)),thereby generating a comparison signal having a duty ratio depending onthe comparison result. In other words, the logic of the comparisonsignal goes to a high level if the error voltage Verr is higher than theslope voltage Vslp and goes to a low level if the error voltage Verr islower than the slope voltage Vslp.

Herer, the on duty (the ratio of on times of the transistors N1, N2 perunit time) of the comparison signal in a steady operation time changesdepending on a relative height difference between the error voltage Verrand the slope voltage Vslp.

During a time the above comparison signal is maintained at the lowlevel, the driver control portion 213 holds the OUTH terminal and theOUTL terminal (i.e., the gate voltages of the transistors N1, N2) at thehigh level via the driver 214 and the driver 216. Accordingly, thetransistors N1, N2 are brought into the on state. On the other hand,during a time the comparison signal is maintained at the low level, theterminal voltages of the OUTH terminal and the OUTL terminal are held atthe low level. Accordingly, the transistors N1, N2 are brought into theoff state.

As described above, in the voltage step-up and down DC/DC controllerblock, the drive control of the transistors N1, N2 is performed based onnot only monitoring results of the LED terminal voltages V1 to V4 (andthe output voltage VOUT) but also monitoring results of the switchcurrent that flows in the transistor N1. Accordingly, in thesemiconductor device 20 according to the present embodiment, even if theerror voltage Verr is not able to follow a sharp change in the load, itis possible to directly perform drive control of the transistors N1, N2based on the monitoring results of the switch current that flows in thetransistors N1, N2; accordingly, it becomes possible to effectively curba change in the output voltage VOUT. In other words, in thesemiconductor device according to the present embodiment, because it isnot necessary to enlarge the capacity of the output capacitor CVOUT andit is possible to use a low-ESR ceramic capacitor, it becomes possibleto avoid an unnecessary cost increase and a size increase in the outputcapacitor CVOUT.

Next, improved points of the protection circuit in the semiconductordevice 20 are described.

First, in the semiconductor device 20 according to the presentembodiment, the type of detecting a short in the open/short detectionportion 222 is changed to a delay counter type. Specifically, astructure is employed, in which the circuit operation is not immediatelymade off-latch at a time any of the LED1 terminal to the LED4 terminalreaches 4.5 [V] but made off-latch at a time it is confirmed that any ofthe LED1 terminal to the LED4 terminal continuously exceeds 4.5 [V] fora predetermined time. By employing such type, it becomes possible toeffectively prevent erroneous detection.

Second, in the semiconductor device 20 according to the presentembodiment, the protection function portion that detects shorts (chieflya ground short) in the anode and cathode of the LED and performssuitable protection operation is incorporated. Specifically, the SCPportion 204 has a structure in which when the SCP 204 portion confirmsthat the terminal voltage VP of the OVP terminal is equal to or lowerthan a predetermined voltage for a predetermined time, the SCP 204recognizes that the anode terminal of the LED train ground-shorts (orshorts to a low potential comparable to this), and off-latches thecircuit operation. Besides, the open/short detection portion 222 has astructure in which when the open/short detection portion 222 confirms byusing an existing open detection function portion that any of the LEDterminal voltages V1 to V4 is equal to or lower than a predeterminedvoltage for a predetermined time, the open/short detection portion 222recognizes that the cathode terminal of the LED train ground-shorts, andoff-latches the circuit operation. By incorporating such protectionfunction portion, it becomes possible to further raise the safety of thesemiconductor device 20.

Third, unlike the first embodiment that turns off the load switch Q1 ata stop time of the circuit operation, the semiconductor device 20according to the present embodiment has a structure in which the OCPsignal and the OVP signal are input into the soft start portion 218, anda soft start voltage (a charging voltage for the capacitor CSS) ispulled down at a time of occurrence of trouble. According to suchstructure, because the soft start is performed again at a time ofresumption of the circuit operation, it becomes possible to prevent arush current at the resumption time.

Finally, an energy-saving function of the semiconductor device 20 isdescribed.

The semiconductor device 20 according to the present embodiment has astructure in which the control logic portion 208 is equipped with atimer latch function; when it is confirmed that the PWM signal ismaintained at the low level for a predetermined time, the control logicportion 208 shifts to an energy-saving mode (sleep mode) for loweringthe consumed energy of the semiconductor device 20. According to suchstructure, it becomes possible to achieve the energy saving of thesemiconductor device 20. Besides, in the above energy-saving mode, it isdesirable not to block the supply route of the power-supply voltage VCCbut to turn off operation of a drive current generation portion (notshown) that generates the drive current ICC for each circuit portion.

Here, in the above embodiments, as application targets of the presentinvention, semiconductor devices that perform the drive control of abacklight of a car navigation monitor, backlights of medium- andsmall-sized LCD panels are described as the examples; however, theapplication target of the present invention is not limited to these, andthe present invention is widely applicable to other load drive devices.

Besides, it is possible to make various modifications to the structureof the present invention without departing from the spirit of thepresent invention.

INDUSTRIAL APPLICABILITY

The present invention is a preferred technology for a drive device thatperforms drive control of a load (a LED backlight of medium- andsmall-sized LCD panels and the like).

1-2. (canceled)
 3. A driving device comprising: an output terminal thatoutputs a pulse signal; a plurality of input terminals connected withone terminal of a plurality of light emitting elements driven by avoltage generated based on the pulse signal; a constant-current driverconnected to each of the plurality of input terminals; a control portionarranged to generate the pulse signal based on a signal which is inputfrom the plurality of input terminals; and an open detection portionconnected with each of the plurality of input terminals and arranged todetermine a connection state of the plurality of light emitting elementsbased on a signal which is input into the plurality of input terminals,and arranged to control the constant-current driver based on a result ofthe determination.
 4. The driving device according to claim 3, whereinthe control portion includes: an error amplifier into which a signalfrom the plurality of input terminals and a reference voltage are inputand which is arranged to amplify and output an error between the signaland the reference voltage; an oscillator; and a comparator arranged togenerate the pulse signal based on an output from the error amplifierand an output from the oscillator.
 5. The driving device according toclaim 3, wherein the constant-current driver includes: a firsttransistor, wherein a pulse-shape current is input into a first terminalof the first transistor, and wherein a second terminal of the firsttransistor is connected with a control terminal; a second transistorincluding a first terminal connected with at least one load, the secondtransistor including a second terminal connected with a referencepotential together with the second terminal of the first transistor, andthe second transistor further including a control terminal connectedwith a control terminal of the first transistor; and a resistor elementconnected between the control terminal of the first transistor and thesecond terminal of the first transistor.
 6. The driving device accordingto claim 5 further comprising a first voltage supply portion and asecond reference voltage supply portion that separately supply apredetermined voltage to one of terminals of the first transistor andone of the terminals of the second transistor.
 7. The driving deviceaccording to claim 3 wherein the open detection portion is arranged todetermine that the plurality of input terminals are opened if a voltageat the plurality of input terminals becomes a predetermined voltage orlower.
 8. The driving device according to claim 7 wherein the opendetection portion is arranged to turn off only a channel connected witha terminal of the plurality of input terminals that is judged to beopened.
 9. The driving device according to claim 3 further comprising: asecond input terminal to receive a voltage based on a voltage suppliedto the light emitting elements; and